Power electronic circuits are used to control and condition electric power. For instance, power electronic circuits may be used to convert a direct current into an alternating current, to change voltage or current magnitude, or to change the frequency of an alternating current.
An inverter is a power electronic circuit which receives a dc source signal and converts it into an ac output signal. Harmonic neutralization and pulse-width modulation techniques are used to generate the ac signal. Harmonic neutralization involves a combination of several phase-shifted square-wave inverters, each switching at the fundamental frequency. Pulse-width modulation involves switching a single inverter at a frequency several times higher than the fundamental.
Inverter switching action generates transients and spurious frequencies in a power signal, usually in the form of harmonics of the switching frequency. The switching action may also produce electromagnetic interference (EMI) which is radiated or conducted through the supply line. While the internal design of an inverter is chosen to minimize transients and spurious frequencies, it is usually necessary to filter the input or the output of the inverter.
Filters can be classified according to whether their main purpose is to improve the power waveform or to remove EMI. Filters for waveform improvement usually deal with frequencies in the audio range. EMI filters are usually concerned with frequencies of 455 kHz or higher.
Passive filters are typically used to eliminate undesirable harmonics from the inverter output. Unfortunately, passive filters do not provide continuous harmonic filtering on pulsating or randomly varying loads. This occurs because passive filters only adapt to new harmonic levels after a considerable settling delay. Passive filters tend to be large, heavy, costly, and, in general, highly load-dependent. Consequently, passive filters frequently represent a substantial part of the total cost, weight, and size of power electronics equipment.
Active filters represent an emerging technology without many of the shortcomings associated with passive filters. The technology relies upon the theory of active-feedback filters. A feedback loop with a single energy-storage element (an inductor or capacitor) is used to minimize the difference between the actual waveform and the desired waveform.
The urgency of developing successful active power filters has recently grown in view of the increasing waveform distortion of both voltages and currents in ac power distribution systems. These distortions are largely attributable to a growing number of nonlinear loads in the electric utility power network. Typical nonlinear loads are computer controlled data processing equipment, numerical controlled machines, variable speed motor drives, robotics, medical and communication equipment.
Nonlinear loads draw square wave or pulse-like currents instead of purely sinusoidal currents drawn by conventional linear loads. As a result, nonlinear current flows through the predominantly inductive source impedance of the electric supply network. Consequently, a non-linear load causes load harmonics and reactive power to flow back into the power source. This results in unacceptable voltage harmonics and load interaction in the electric power distribution in spite of the existence of voltage regulators.
The degree of current or voltage distortion can be expressed in terms of the relative magnitudes of harmonics in the waveforms. Total Harmonic Distortion (THD) is one of the accepted standards for measuring voltage or current quality in the electric power industry.
Apart from voltage and current distortion, another related problem may arise when nonlinear loads are connected to the electric power network. In particular, when the load current contains large amounts of third or other triplen harmonics, the harmonic current tends to flow in the neutral conductor of the power system. Under these conditions, the neutral current can exceed the rated current of the neutral conductor. Since the neutral is normally designed to carry only a fraction of the line current, overheating or even electric fires can result.
As previously indicated, active filters may be used to alleviate these problems. Active filters, or active power line conditioners (APLCs); comprise one or two pulse width modulated inverters in a series, parallel, or series-parallel configuration. The inverters share a common dc link, which can be a dc inductor (current link) or a dc capacitor (voltage link). It is advantageous to keep the energy stored in the dc link (capacitor voltage or inductor current) at an essentially constant value. The voltage on the dc link capacitor can be regulated by injecting a small amount of real current into the dc link. The injected current covers the switching and conduction losses inside the APLC. The link voltage control can be performed by the parallel inverter.
The basic active load current compensation with current or voltage source filters is known. FIG. 1 depicts a parallel connected current source active filter 20, and FIG. 2 depicts a parallel connected voltage source active filter 22. The load current I.sub.L consists of three components: The real current, I.sub.r, the reactive current, I.sub.q, and the ripple current, I.sub.R. The parallel connected active filter supplies the I.sub.R and I.sub.q components, and, also, a small residual "high frequency" component I.sub.hf, that flows into the parallel connected "high frequency" capacitor C.sub.hf. The parallel connected active filter is essentially a single or multi-phase inverter which is operated from an isolated current or voltage source.
The realization of the active filter requires solid state switches with intrinsic turn-off capability (transistors, IGBTs, MOSFETs, GTOs, etc.). Switch pairs P1 and P2 are alternately turned ON and/or OFF. The average voltage required in the link capacitor, V.sub.dc, of FIG. 2, is supplied by the ac source. Real power can be absorbed by introducing an appropriate amount of offset in the symmetry of the on-times in switches P1 and P2. The polarity of the offset is coordinated with the polarity of the input voltage. When switches P1 of FIG. 2 are on, a resonant current is generated between the tie inductor, Lp, the output capacitance dominated by C.sub.hf, and the difference between the dc link and ac output voltages. Conversely, when the P2 switch pair is on, the resonant current is driven by the sum of the dc link and ac output voltages. Since the dc link voltage is regulated to be larger than the peak value of the ac voltage, the voltage polarity that drives the resonant current will reverse after each complementary pole switching.
The real power, necessary to maintain the selected dc link voltage magnitude, Vdc, is proportional to the average duty cycle of high-frequency pole switchings in any given half line voltage cycle. The isolated dc link voltage is regulated by a closed loop controller that affects the average pole switching symmetry. Reactive inverter currents can be produced that flow in or out of the inverter by temporary changes in the duty cycle of inverter pole switchings. The instantaneous magnitudes of inverter currents are regulated so that they provide the load compensation current requirements. For example, if a positive ripple current is detected, the on-time of P2 is increased with respect to P1. The increase results in the required net compensating ripple current flowing in the ac line. This also implies that the amplitude of Vdc must be kept higher than the highest value of the ac voltage across the load, otherwise, the instantaneous compensation capability of the active filter is impaired.
The rapid pulse width modulation switching in the active filter produces a high frequency, typically, triangular shaped current, I.sub.hf, an undesired side effect. The effect of the I.sub.hf signal is a small, superimposed saw-tooth voltage ripple on the ac voltage. With a given tie inductor value, the amplitude of the voltage ripple is inversely proportional to the pole switching (carrier) frequency and the value of C.sub.hf. The voltage ripple is filtered with a parallel capacitor C.sub.hf.
When the active power filter (20 or 22) is connected across the load, a high degree of filtering of the terminal voltage is observed. Note that the active power filter is not capable of supplying or absorbing any real power other than that which is needed to compensate for losses inside the filter itself. It will, however, readily compensate reactive currents, non-synchronous and non-theoretical harmonics and sources with variable or unregulated frequency. The shunt connected power circuit is inherently protected under load short circuits since the load fault current bypasses the active power filter.
The isolated dc link circuits of FIGS. 1 and 2 can be combined to produce an ac line conditioner and voltage regulator. FIG. 3 depicts a shared link current source active power filter 24 with a serial inverter 26 and a parallel inverter 28. FIG. 4 depicts a shared link voltage source active power filter 30, with a serial inverter 32, and a parallel inverter 34. The respective series and parallel inverters are similar to the filters described in relation to FIGS. 1 and 2. The shared link approach of FIGS. 3 and 4 represents a combination of series and shunt connected filters which are operated from a common shared direct voltage (or current) source.
The shared link circuit topology removes the former limitation of the active power filter, namely, that it is not capable of supplying or absorbing any real power, apart from compensating for the losses in the active power filter itself. In the shared dc link series and parallel circuit topology, it becomes possible for both the series and the parallel filter elements to absorb or generate real power at the fundamental frequency, or other frequencies, provided the total power absorbed equals the total power generated.
The series active elements (26 and 32) may be modulated to provide a fundamental voltage of controllable magnitude and phase so that the phase and magnitude of the ac output voltage stays sinusoidal at any required level and phase angle with respect to the ac input. The power required by the series element (26 or 32) is absorbed from or injected into the dc link (36 or 38). Link energy is then maintained by appropriately controlling the phase and magnitude of the fundamental modulating signal applied to the parallel connected element (28 or 34). The result is that the power needed by the series element (26 or 32) will be obtained from the parallel element (28 or 34). Similarly, power generated by the series element (26 or 32) will be returned into the ac output by the parallel element (28 or 34).
When the output and input voltages are not equal, the series inverter (26 or 32) delivers real power to or from the dc link (26 or 38). The amount of power exchange delivered with respect to the output power depends on the fundamental Vo/Vin ratio. When the Vo/Vin ratio is smaller than unity, the real part of the input current becomes larger than the output (load) real current. The difference between the output and input currents flows through both inverters via the dc link (36 or 38). Appropriate fast-acting controls insure that the power flow between the series and parallel inverters is kept equal on the average, so that the power flow does not significantly alter the stored energy in the shared dc link.
In addition to the regulation of the buck/boost power transfer, the parallel active element (28 or 34) is modulated at ripple frequency so that it provides a bypass for load generated ripple currents and, if required, for the reactive fundamental current of the load. After full compensation of ripple and reactive components, only real fundamental current is drawn from the ac input.
In spite of sharing a common dc link, both the series and parallel inverters can be independently controlled to exhibit reactance at fundamental frequency, by appropriate adjustments of their fundamental modulating signals. This is useful for phase correction of the output voltage if necessary, and for compensating unbalanced reactive loads in multiphase ac power systems. Another important observation is that due to the nature of the active power filter, its operation as a variable capacitor at fundamental frequency will not cause any undesirable resonances at harmonic or modulation frequencies.
Certain types of electrical loads, such as synchronous and induction motors, require balanced three phase voltages. Small voltage unbalances in such devices, attributable to a negative sequence fundamental, can result in significantly larger current unbalances, resulting in an over-current stator winding condition, excessive stator winding temperature, excessive motor noise, and higher motor core losses. Thus, motor lifetime and reliability are adversely affected. Therefore, a desirable Active Power Line Conditioner will have the capability to provide balanced load voltages in the presence of unbalanced source voltages. It is also desirable to have the capability to force balanced source currents if the load currents are unbalanced due to unbalanced phase impedances. In other words, it is desirable to provide an active power line conditioner which eliminates the negative sequence fundamental.
Active power line conditioners endeavor to improve the power available for a load. Harmonics in a load current reduce the power factor of the load, while causing distortion and interference problems. Anything less than unity power factor reduces the amount of power that can be drawn from an ac power supply. The reduction of available power is characterized by the given line impedance of the electric network. Input power factor (PF) is defined as the ratio of true input power, Pt, to apparent input power, Pa: PF=Pt/Pa. If the ac input voltage is sinusoidal, the true input power is the result of the rms value for the input voltage multiplied by the rms value of the in-phase component of the fundamental input current. The "in-phase" component of the fundamental input current is a sinusoidal quantity having a zero phase displacement with respect to the sinusoidal input voltage. The harmonic currents of the load increase the rms value for the load current but not the fundamental value. Only the in-phase component of the fundamental load current contributes to the power consumed by the load. Generally, the phase of the load current fundamental is not in phase with the input voltage. If the input voltage is not sinusoidal, the calculation of the load power factor becomes even more complex.
Equipment with nonlinear load currents may include conventional linear lagging or leading loads. The application of linear lagging or leading loads results in less than unity "displacement power factor". The conventional displacement power factor, DPF, is caused by phase shift in the fundamental as opposed to the load harmonics caused power factor, HPF, that appears as an increase in the rms value of non-sinusoidal load current. Conventional loads are, for example, cooling fans with lagging power factor. These two kinds of power factors can coexist in varying ratios but each of them affects the terminal voltage in an adverse fashion and, therefore, some of the other loads connected to the power bus.
Thus, a unified power factor controller is needed to alleviate power factor related efficiency and load voltage drop problems at the input terminals of an active power line conditioner. It would be highly desirable to provide a unified power factor controller with fast dynamic response following step changes caused by the application or removal of a load.
The series transformer of an active power line conditioner, by default of its series connection, must carry the difference of currents between the active power line conditioner load and the parallel inverter. The inrush current problem into the dc link capacitor on start-up of the APLC is generally solved by dedicated link current limiting circuitry. The handling of load current surges without undue overrating of APLC components is an important consideration in producing a cost-effective commercial product.
In relatively smaller power applications, the series transformer has a higher than 1:1 turns ratio. The selected higher than 1:1 ratio is in accordance with the particular series voltage support requirement for the series inverter. When this is the case, the series inverter is not capable of full limitation of the short circuit current. The magnitude of the short circuit current will be determined by the unsupported line voltage and source/short-circuit impedances in the overall network. On the other hand, the non-unity transformer ratio means higher reflected ac voltages that are applied to the series inverter. If the series inverter is rated to handle the high surge voltages, the parallel inverter must also be rated to the same high voltage, since the two inverters share a common dc voltage link. In fact, the dc link must be charged to a higher than peak ac voltage level in order to maintain current control and thus avoid false, series inverter over-current trips. To rate the inverters for these surge voltage and surge current rating requirements may not result in a commercially competitive product. Thus, it is important to develop a cost-effective APLC which complies with surge rating requirements in a different way.
The surge protective functions override the active power quality controllers in an active power line conditioner. Consequently, a protective function, while in effect, can result in a temporary compromise in the output power quality, such as: elimination of output voltage regulation, injection of load harmonics back into the source, and uncompensated input voltage harmonics. Thus, it would be highly desirable to provide an active power line conditioner which quickly recovers from transient conditions when internal protections are activated. In other words, it would be highly desirable to provide an active power line conditioner which is highly tolerant to system transients and to the activation and deactivation of active power line conditioner protective features.
A number of control strategies are applied to ac machines. In general, ac machine control theory is directed toward providing accurate mechanisms for controlling the torque of a machine. Torque control in an ac machine is obtained by managing a current vector composing amplitude and phase terms. The control of ac machines is complicated by the requirement of external control of the field flux and armature mmf spatial orientation. In the absence of such a control mechanism, the space angles between the various fields in an ac machine vary with load and result in oscillations or other unfavorable physical phenomenon. Control systems for ac machines which directly control the field flux and armature mmf spatial orientation are commonly referred to as "field orientation" or "angle" controllers. Such controllers employ synchronous transformations, as will be described below.
The fundamental principles of field orientation control of ac motors is described in Introduction to Field Orientation and High Performance AC Drives, IEEE Industrial Drives Committee of the IEEE Industry Applications Society, Oct. 6-7, 1986. Field orientation principles rely upon the fact that the rotor of a motor has two axes of magnetic symmetry. One axis is known as the direct axis, and the other axis is known as the quadrature axis. These terms are usually shortened to simply refer to the d-axis and the q-axis.
Field orientation techniques endeavor to control the phase of the stator current to maintain the same orientation of the stator mmf vector relative to the field winding in the d-axis within the d-q scheme. FIG. 5 depicts a symbolic representation of a field orientation control system and its corresponding mathematical model. The three phase system (a, b, c) is first synchronously transformed to a two phase ds-qs scheme which is stationary with respect to the three phase system. This 3-phase to 2-phase transformation is equivalent to a set of linear equations with constant coefficients, as shown in FIG. 5.
The second step is the synchronous transformation from stationary d-q variables to rotating d-q variables. This transformation involves the angle 8 between the two systems and is described by the matrices given in the figure. The rotation transformation is often referred to as a "vector rotation" since the d-q quantities can be combined as a vector and the transformation then amounts to the rotation of one vector with respect to the other. FIG. 5 includes the vector rotation equations.
FIG. 6 depicts the inverse synchronous transformations to those performed in FIG. 5. Initially, a rotating-to-stationary synchronous transformation is made using the matrices depicted in FIG. 6. After the stationary rotor reference frame variables are established, a two phase to three phase synchronous transformation is made, consistent with the equations provided in the figure.